1. Field of the Invention
This invention relates to an isolated dc/dc converter. More particularly, this invention relates to a constant-frequency, isolated dc/dc full-bridge converter that operates with zero-voltage switching of the primary switches.
2. Discussion of the Related Art
A factor adversely affecting the performance of a conventional xe2x80x9chard-switchedxe2x80x9d pulse-width-modulated (PWM) converter at a high switching frequency is circuit parasitics, such as semiconductor junction capacitances, transformer leakage inductances, and rectifier reverse recovery. Generally, these parasitics introduce additional switching losses and increase component stresses, thus limiting the converter""s maximum operation frequency. To operate a converter at a high switching frequency and to achieve a high power density, elimination or a reduction of circuit parasitics without degrading conversion efficiency is required. One approach which incorporates circuit parasitics into circuit operations uses a resonant technique or a constant-frequency PWM soft-switching technique.
Under a resonant technique, a resonant tank circuit shapes the current or voltage waveforms of semiconductor switches in the converter to create either zero-current turn-off, or zero-voltage turn-on conditions. However, relative to conventional switching techniques, zero-current switching (ZCS) and zero-voltage switching (ZVS) in a resonant-type converter cause higher current or voltage stresses in the semiconductor switches. In addition, to create a ZCS or a ZVS condition, a resonant topology typically circulates a significant amount of energy. Thus, the trade-off between switching loss and conduction loss may result in a lower efficiency or a larger high-frequency resonant-type converter, when compared to a PWM counterpart operating at a lower frequency, especially in an application involving a wide input voltage range. In addition, variable frequency operation is often seen as a disadvantage of resonant converters. As a result, although resonant converters are used in a number of niche applications, such as those with pronounced parasitics, the resonant technique has not gained wide acceptance in the power supply industry in high-frequency, high-power-density applications.
To overcome the degradation of efficieny due to circuit parasitics, a number of techniques that enable a constant-frequency PWM converter to operate with ZVS or ZCS have been proposed. In such a soft-switching PWM converterxe2x80x94one that possess the PWM-like square current and voltage waveformsxe2x80x94lossless turn-off or turn-on of the switches can be achieved without a significant increase of conduction loss. One soft-switched PWM circuit is the soft-switched, full-bridge (FB) PWM converter 100 of FIG. 1(a), which is discussed in the article xe2x80x9cPseudo-Resonant Full Bridge DC/DC Converter,xe2x80x9d by O. D. Petterson, D. M. Divan, published in IEEE Power Electronics Specialists"" Conf: Rec., pp. 424-430, 1987, and in the article xe2x80x9cDesign Considerations for High-Voltage High-Power Full-Bridge Zero-Voltage-switched PWM Converter,xe2x80x9d by J. Sabate, et. al., published in IEEE Applied Power Electronics Conf: (APEC) Proc., pp. 275-284, 1990. Converter 100 provides ZVS in the primary switches with relatively small circulating energy and at a constant switching frequency. A constant-frequency output voltage is achieved by a phase-shift technique. Under this technique, a switch in the lagging leg (i.e., switches 103 and 104) of the bridge is closed only after a delay (i.e., phase shifted) relative to the closing of a corresponding switch in the leading leg (i.e., switches 101 and 102), as shown in FIG. 1(b). Without the phase-shift, no voltage is applied across the primary winding 105a of transformer 105, resulting in a zero output voltage. However, if the phase-shift is 180xc2x0, the maximum volt-second product is applied across the primary winding 105a, which produces a maximum output voltage. In converter 100 of FIG. 1(a), a ZVS condition in the lagging-leg (i.e., switches 103 and 104) is achieved by the energy stored in output filter inductor 106. Since filter inductor 106 is relatively large, the energy stored in filter inductor 106 is sufficient to discharge output parasitic capacitances 107 and 108 of switches 103 and 104 to achieve the ZVS condition, even at a small load current. However, parasitic capacitances 112 and 113 of leading-leg switches 101 and 102 are discharged by energy stored in leakage inductance 109 of transformer 105. (During the switching of switches 101 and 102, primary winding 105a is shorted by rectifiers 110 and 111 carrying the output current of filter inductor 106.) Since leakage inductance 109 is small, switches 101 and 102 cannot achieve ZVS condition even at relatively high output currents. In the prior art, the ZVS range of leading-leg switches 101 and 102 is extended either by increasing leakage inductance 109, or by adding an external inductor in series with primary winding 105a. A properly sized external inductor can store enough energy to achieve ZVS condition in the leading-leg switches 101 and 102 even at low currents. However, a large external inductor would also store a large amount of energy at the full load, thus producing a large circulating energy adversely stressing the semiconductor components and reducing conversion efficiency. Further, in converter 100, severe parasitic ringing may occur in the secondary winding 105b of transformer 105 when one of rectifiers 110 and 111 turns off. Such ringing results from a resonance among the junction capacitance of the rectifier, leakage inductance 109 and the external inductor (when present). To control such ringing, a snubber circuit is required on the secondary side of transformer 105, thus significantly lowering the conversion efficiency of the circuit.
Alternatively, in the prior art, the ZVS range of switches 101 and 102 is extended to lower load currents without a significant increase of the circulating energy by using a saturable external inductor, as illustrated by full-bridge ZVS PWM converter 200 of FIG. 2. (In this discussion and in the detailed description below, to facilitate correspondence between figures, like elements are assigned like reference numerals). Converter 200 is described in the article, xe2x80x9cAn Improved Full-Bridge Zero-Voltage-Switched PWM Converter Using a Saturable Inductor,xe2x80x9d by G. Hua, F. C. Lee, M. M. Jovanovic, published IEEE Power Electronics Specialists"" Conf: Rec., pp. 189-194, 1991. When saturable inductor 209 is sufficiently large to saturate at a high load current, a controlled amount of energy is stored in saturable inductor 209. At the same time, at a low load current (i.e., when saturable inductor 209 is not saturated), saturable inductor 209 has a sufficiently high inductance to store enough energy to provide ZVS in switches 101 and 102 even at small loads. However, when placed in the primary side of transformer 201, saturable inductor 209 requires a relatively large magnetic core, thus increasing the cost of converter 200. (Generally, a large magnetic core is required to eliminate excessive heat resulting from core loss as the flux in a saturable inductor swings between the positive and negative saturation levels).
In the prior art, the ZVS range of a FB ZVS PWM converter is also extended to a lower load current by placing saturable inductors on the secondary side, as illustrated by FB ZVS PWM converter 300 of FIG. 3. As shown in FIG. 3, saturable inductors 309a and 309b are connected in series with rectifiers 110 and 111, so that the flux swing in each of saturable inductors 309a and 309b is confined between zero and a positive saturation level (i.e., the flux swing in each of saturable inductors 309a and 309b is approximately half the flux swing of saturable core 209 of FIG. 2.) As a result, core loss in converter 300 in FIG. 3 is reduced, as compared to converter 200 of FIG. 2. However, because in voltage step-down converters (i.e., converters with an output voltage Vo smaller than input voltage Vin) secondary currents are larger than the primary current, the total copper loss of the windings of saturable inductors 309a and 309b is increased, when compared to the copper loss of the windings in saturable inductor 209. Secondary-side saturable inductors 309a and 309b serve as turn-off snubbers for rectifiers 110 and 111, thus damping the parasitic oscillations between the junction capacitance of rectifiers 110 and 111 and the leakage inductance of transformer 301, and reducing the reverse-recovery current losses when fast-recovery rectifiers are used.
In a FB ZVS PWM converter with secondary-side saturable inductors, such as converter 300, a freewheeling rectifier 302 may be used. With freewheeling diode 302, saturable inductors 309a and 309b store enough energy at lower load currents so that a ZVS condition for the primary switches is achieved with minimum circulating energy. Without freewheeling diode 302, saturable inductors 309a and 309b are not used for energy storage, as explained in U.S. Pat. No. 5,132,889,xe2x80x9cResonant-Transition DC-to-DC Converter,xe2x80x9d to L. J. Hitchcock, M. M. Walters, R. A. Wunderlich, issued on Jul. 21, 1992. Instead, saturable inductors 309a and 309b are used to briefly delay turning on the non-conducting one of rectifiers 110 and 111 after a corresponding switch in a bridge leg is opened, so that the current in filter inductor 106 continues to flow through the conducting one of rectifiers 110 and 111. As a result, in converter 300, the energy stored in filter inductor 106 creates a ZVS condition for switches 101 and 102 in the same way as it creates a ZVS condition for switches 103 and 104.
Finally, in a FB ZVS PWM converter, any inductance connected directly in series with the primary or secondary winding (or both) including the leakage inductance of the transformer, causes a loss of duty cycle at the secondary side of the transformer. The loss of duty cycle is detrimental to efficiency, since a lower duty cycle requires a reduced number of turns in the transformer, which increases both conduction loss in the primary side and voltage stresses in components of the secondary side. The loss of duty cycle results from the commutation time required for the primary current to change direction. Because, during the commutation time, the windings of the transformer are shorted by all the secondary side rectifiers simultaneously conducting, the commutation time, and therefore the duty cycle loss, is proportional to the total inductance connected in series with the transformer windings.
Because circuit 100 in FIG. 1 requires an increased leakage inductor or an external inductance (or both) in series with the transformer for ZVS, circuit 100 suffers from a large loss of duty cycle on the secondary side. Converter 200 of FIG. 2 and converter 300 of FIG. 3 have a smaller duty cycle loss, since they use saturable inductances, which reduces the effective commutation inductance. Generally, the optimal FB ZVS PWM converter should be able to achieve ZVS of primary switches without a need for external linear or saturable inductors, and with a minimum leakage inductance (preferably zero).
The present invention provides an isolated, constant-frequency, dc/dc FB ZVS PWM converter which employs a coupled inductor on the primary side of the transformer to achieve a ZVS condition for the switches in the full bridge over wide ranges of load currents and input voltages. A converter of the present invention has reduced circulating energy and conduction losses. In one embodiment, two windings of a coupled inductor are connected in series and their common terminal is connected to one end of the primary winding of a transformer (the other end of the primary winding is connected to ground). The other terminals of the coupled inductor are respectively connected to midpoints of two bridge legs through a corresponding blocking capacitor. The secondary side of such a converter can be implemented using a full-wave rectifier, such as a full-wave rectifier with a center-tap secondary winding, a full-wave rectifier with a current doubler, or a full-bridge full-wave rectifier. The output voltage regulation in the converter is achieved by employing a constant-frequency phase-shift control.
In a converter of the present invention, both the energy stored in an output filter inductor and the magnetizing inductance of the coupled inductor are used to discharge the parasitic capacitance across a switch to achieve a ZVS condition. Since the coupled inductor transfers current (hence, energy) from the winding in one bridge leg to the other bridge leg, a converter of the present invention opens all bridge switches when the switches carry currents of the same magnitude. As a result, the energy available for discharging the capacitances of each switch is the same for all primary switches.
According to another aspect of the present invention, a converter achieves ZVS conditions for all the primary switches, even in the absence of a load, by properly selecting a value for the magnetizing inductance of the coupled inductor. In a converter of the present invention, because energy is not stored in a leakage inductance, the transformer""s leakage inductance can be minimized, thus significantly reducing the secondary-side ringing caused by a resonance between the leakage inductance and a junction capacitance of the rectifier. Power dissipation in a snubber circuit usually required to damp ringing is also reduced. Moreover, due to a minimized leakage inductance of the transformer, duty cycle loss on the secondary side of the transformer is also minimized. As a result, a converter of the present invention can operate with a very high duty cycle, thus minimizing both conduction loss in the primary switches and voltage stresses on the components of the secondary side, and achieving improved efficiency.
The present invention is better understood upon consideration of the detailed description below and the accompanying drawings.